1 Introduction

It appears that the upward of wireless communication technologies will carry on and their numerous applications have been strictly come to be beyond belief. We have come by different communication technologies starting from 1G, 2G, 3G, long-term evolution advanced (4G LTE-A) up to 5G technologies [1,2,3,4], UWB technology [5,6,7], and satellite communications [4]. One of the inciting issues that affect the current progress in wireless communication is the deficiency of available frequency resources. Accordingly, researches have been initiated in 5G wireless communications at the millimeter frequency bands starting from 6 up to 100 GHz. As a whole, the frequency ranges used for 5G researches are started from 4 to 60 GHz [8, 9]. Several applications have employed the 5G technology with the Internet of Things (IOT). The great challenge of 5G wireless communications is to join millions of cellular devices collectively as in modern cities and transportation. Reduction in the dimension of cellular phones depends on the improvement of minute antennas embedded in these devices. Thus, the raise of microstrip antennas designs has curricular roles in the twentieth century. Generally, the microstrip patch antennas are arranged by a thin metal layer mounted on a dielectric substrate backed with thin metal sheet [10]. The microstrip patch antennas are easily integrated on PCB's surface for 5G cellular devices. Typically, these antennas are used in multifrequency bands with several advantageous features such as lightweight, low cost, and conformal to either planar or nonplanar surfaces.

Researchers offered different antenna types to cover the different planned 5G frequency bands.

The most common types of antennas are target-specific, multiband, and frequency reconfigurable antennas. Also, these approaches have adopted many techniques like metamaterials, defected ground structure (DGS) to achieve wideband microstrip antennas for 5G applications [11]. In [12], the UWB performance has been achieved with a microstrip antenna that resonates at four discrete frequencies: 3.2, 5.1, 6.9, and 8.3 GHz, respectively. Two applied cases of single patch antennas through the material of a dielectric permittivity constant equal to 2.43 are considered in [13]. The prototypes of the two built antennas provided resonances at 32 and 62 GHz. A low-cost printed circuit board (PCB) based a dual band antenna for wireless communications is presented in [14]. The antenna is mainly designed for both Wi-Fi and Wireless Gigabit Alliance (WiGig) applications. The results show good impedance bandwidth across the Wi-Fi bands, however, at WiGig band; the fixture has little effect on the performance of the antenna. In [15], a circularly polarized well-matched square microstrip patch antenna is offered. The offered antenna has two wide bandwidths extend from 21 to 62 GHz with VSWR values nearly 2. In this study, an antenna design has been achieved which enables the operation of two multiband from 25:32 and 40:60 GHz. Its rectangular and circular patches are loaded on a single square-shaped microstrip patch antenna (MPA) structure with a planar transmission line (T.L) feed. The significant features of the proposed antenna can be summarized as follows: (1) Simple rectangular and circular patches are loaded on a single square-shaped microstrip patch for shifting frequency to lower and upper bands without aggregate the antenna size; (2) it reveals two wide operational bandwidths of simple geometrical configuration without any defected ground structures (DGSs) that were used in the reported literature; (3) the combinations of compact size, high gain, and almost constant radiation efficiency over a wide range of frequencies recommend a future usage of the offered antenna in the modern 5G communication applications. As a result, this antenna was able to be operated in multi-band with antenna efficiency reaching up to 84% and up to 10.75 dB gain values.

The paper is divided as follows: Section 2 describes the antenna design and working principle. In Section 3, simulations results of the engineered structures with different techniques have been offered. Fabrication and tests are then presented in Section 4; finally, Section 5 presents the conclusion.

2 Antenna Design and Working Principle

The offered rectangular-circular shaped microstrip antenna is principally planned to cover the communication requests that operate at both 25–32 and 40–60 GHz frequency ranges. The whole antenna structure is embedded on a single low loss substrate layer (Duroid-Rogers RO4003) directly above the ground plane. Its dielectric thickness is 0.203 mm, dielectric constant of 3.55, and loss tangent of 0.0027. The engineered antenna is then based on the design of four radiator elements, two rectangular and two circular patches. Thus, different resonance frequencies at different bands can be fulfilled. The four patches are excited directly by a microstrip line through a tapered inset feed. In relation to the elementary formulas present in [16], both rectangular and circular patches can be designed, wherein the analysis of the proposed antenna rectangular patch starts by calculating the conventional width (\(\mathrm{L}\)) and length (\(\mathrm{W}\)) by the following formulas [16, 17]:

$$L=\frac{{\varvec{C}}}{{2 {\varvec{F}}}_{{\varvec{r}}}}\sqrt{\frac{2}{{\in }_{{\varvec{r}}}+1}}$$
(1)

The rectangular patch effective length and the substrate effective dielectric constant will be calculated respectively as follows:

$$W=\frac{1}{{2 F}_{r} \sqrt{{\in }_{eff}} \sqrt{{\epsilon }_{0}{\mu }_{0}} }-2\nabla \mathrm{W}$$
(2)

and

$${\in }_{{\varvec{e}}{\varvec{f}}{\varvec{f}}}=\frac{{\in }_{r}+1}{2}+\frac{{\in }_{r}-1}{2} {\left(1+12 \frac{h}{L}\right)}^{-0.5}$$
(3)

where \({F}_{{\varvec{r}}}\) is the desired resonant frequency, \({\in }_{{\varvec{r}}}\) is the relative dielectric constant of the substrate, and \(h\) represents the thickness of the antenna substrate which is equal to 0.203 mm in the proposed design.

\(\nabla \mathrm{W}\) is the normalized extension in the rectangular patch length, and it will be mathematically specified as mentioned in [16, 17] by:

$$\nabla \mathrm{L}=0.412h* \left\{\left(\frac{{\in }_{{\varvec{e}}{\varvec{f}}{\varvec{f}}}+0.3}{{\in }_{{\varvec{e}}{\varvec{f}}{\varvec{f}}}-0.258} \right)*\left(\frac{\frac{{\varvec{L}}}{{\varvec{h}}}+0.264}{{\in }_{{\varvec{e}}{\varvec{f}}{\varvec{f}}}+0.8}\right)\right\}$$
(4)

Similarly, the radius of each circular patch has been designed with the aid of the following equations:

$$R=T{\left[1+\frac{2h}{\pi T {\in }_{r}} \left(\mathrm{ln}\left( \frac{\pi T}{2h}\right)+1.7726\right)\right]}^{-0.5}$$
(5)

and

$$T=(8.719*{10}^{9} )/{(\sqrt{{\in }_{r}} *F}_{rc} )$$
(6)

where \({F}_{rc}\) is the circular lower cut-off frequency in GHz. Originally, it is necessary to design a single patch antenna element when the multiple antenna elements (i.e., an antenna array) are the ultimate objective. Therefore, Eqs. (2–6) are used to find the length and width of the proposed antenna rectangular and circular patches that attain resonances at the desired frequency ranges. We set the initial dimension values for the rectangular and circular patches regarding the above design equations. According to the elementary formulas presented above, both rectangular and circular patch design parameters are introduced in Table 1 [18]. Then, the antenna parameters have been optimized via the finite element method through the ANSYS HFSS software to obtain the best values and then produce a matched antenna around the required operating frequency bands. Consequently, the geometry of the proposed dual-band microstrip patch antenna with optimum design is illustrated in Fig. 1, wherein the top view demonstrates a more detailed perspective of each designed part. Starting with the ground plane is then moving on to the coaxial connector close to the feed line and ended up at different microstrip patches. Also, Table 2 illustrates the decision space for each parameter to be optimized and the optimum values for the proposed antenna parameters. Therefore, the top view of the engineered antenna as shown in Fig. 1 is with an overall area of (L = 24 × W = 24)\({\mathrm{mm}}^{2}\). The double rectangular patches are taking part in the lower and upper antenna arms, each with an inner circular slot. Small slots were embedded at the center of each rectangular patch of radius R1 and R4 by using the optimum parameters to improve the antenna structure performance and enlarge the operating impedance bandwidth [18]. A circular patch of radius R2 is connected to the upper antenna arm, while the lower arm has a circular patch of radius R3. The length and width of the tapered coupled inset feed line are also optimized for the purpose of better matching to the incident wave. To prevent the existence of higher order modes and agreement of 50 Ω microstrip line matching impedance, the attachment metallic feed line cut-off operating frequency will be given by the following [17]:

$${\mathrm{F}}_{\mathrm{L}}=\frac{\mathrm{c}}{{{\in }_{\mathrm{r}}}^{0.5}(2{\mathrm{L}}_{\mathrm{f}1}+\mathrm{W}5)}$$
(7)

where c = \(3*{10}^{8}\) m/sec represents the speed of light and \({\in }_{\mathrm{r}}\) and \({\mathrm{F}}_{\mathrm{L}}\) are the substrate dielectric constant and the feed line frequency, respectively. Also, the estimation of the planned antenna feed line length was performed by using the next characteristic impedance equation:

$${\mathrm{Z}}_{\mathrm{O}}=120\uppi / \left\{\sqrt{{\in }_{\mathrm{eff}}}\left[\frac{{\mathrm{L}}_{\mathrm{f}1}}{\mathrm{h}}+1.393+0.667\mathrm{Ln}\left(\frac{{\mathrm{L}}_{\mathrm{f}1}}{\mathrm{h}}+1.444\right)\right]\right\}$$
(8)

where \({\mathrm{Z}}_{\mathrm{O}}\) is the 50 Ω microstrip line and [17]

Table 1 Antenna design and optimized parameters in (mm)
Full size table
Fig. 1
figure 1

Proposed antenna structure with the radiator patches (top view)

Full size image
Table 2 Simulated and measured radiation patterns 2- D radiation characteristics at both E & H planes for xz plane and yz plane of the proposed antenna: simulated ( ––- Dashed line) and measured ( ______ Solid line) for four different frequencies (a) 25, (b) 30, (c) 52.6 GHz and (d) 54 GHz
Full size table
$$\frac{{\mathrm{L}}_{\mathrm{f}1}}{\mathrm{h}}>1$$

The proper choice of the substrate’s width for obtaining a 50 Ω microstrip line is (\({L}_{f1}\) = 0.4) mm2 [17]. As the reflection of incident wave is reduced based on impedance matching approach, a tapered microstrip feed line with optimized dimensions (W3, W4,) and ( \({L}_{f2},{L}_{f3}\)) mm2 is then made on the same substrate for proper signal excitation. Accordingly, the assigned substrate parameters achieve up to 60 GHz operating frequencies without higher order modes of operation [17]. Finally, the designed patch impedance \({{\varvec{Z}}}_{{\varvec{l}}}\) is calculated by means of the following Equation [16, 17]:

$${Z}_{l}=90*\frac{{{\in }_{{\varvec{r}}}}^{2}}{{\in }_{{\varvec{r}}}-1} {\left( \frac{{\varvec{W}}}{{\varvec{L}}}\right)}^{2}$$
(9)

Consequently, two rectangular and circular patches are loaded on a single square-shaped microstrip patch antenna structure of a planar a 50 Ω feed line as depicted in Fig. 1.

3 Simulations Results

In the current work, our main objective is to exploit the performance for circular and rectangular-shaped patches in the proposed design to operate in the desired bandwidths. The physical parameters (length and width) for the rectangular patches are optimized to make their physical dimensions approximately equal λg/4 while the influence of the circular patches dimensions with almost λg/2. The rectangular and circular loaded radiators are then providing dual wide bandwidths with multiple frequency resonances as shown in Fig. 2. Therefore, the spectrums achieved the ITUrequirements available for 5G mobile communications under the proposed millimeter wave (mm-wave) band [19]. Furthermore, HFSS and CST (\({\mathrm{S}}_{11}\)) results in the two bandwidths depicted in Fig. 2 provide perfect selection for upcoming 5G mobile requests. A higher (\({\mathrm{S}}_{11}\)) power greater than − 20 dB has been fulfilled consecutively from both simulators at different frequency values in the first bandwidth as shown in Fig. 2a, while in Fig. 2b, resonance frequency values ≥ – 10 dB have been fulfilled within the entire bandwidth resulted in the best matching impedance. It should be noted that (\({S}_{11}\)) power values are lesser than – 10 dB for all resonance frequencies within the dual bands which resolve the few critical restrictions encountered by these dual (mm-wave) bands, such as the atmospheric absorptions and signal fading [20]. Considering \({S}_{11}\) characteristics as shown in Fig. 2, the results point out a reasonable agreement between both HFSS and CST \({S}_{11}\) results either in 25–32 or 40–60 GHz bandwidths. The proposed antenna design was performed through two numerical simulation software packages: the HFSS, which uses the finite element method (FEM), and the CST, which uses the finite integral method (FIT). As the proposed antenna operates in a multiband regime, more than a single frequency sweep has been used specially by the HFSS to solve the result. It should be noted that there are differences in the S11 location of notches and their values between the HFSS and CST over the entire band. Consequently, these variations are due to the different numerical techniques along with their operating environments, such as different geometry mesh algorithms, which result in different convergences of the software packages within the two operating wide bandwidths [21].

Fig. 2
figure 2

Simulated \({\mathrm{S}}_{11}\) of the proposed antenna

Full size image

3.1 Surface Current Distribution on the Proposed Antenna Structure

For more explanation of the planned antenna performance, the distribution for the current on the surface of the antenna structure was fulfilled at different frequencies by the HFSS simulator. The anticipated patch loaded radiator’s current distributions at different frequency values are illustrated in Fig. 3, where the highest tone (red color) represents the strongest surface current distribution area. As the main radiation mechanism is due to the surface current distribution around the antenna feed line and on the inner edges of the rectangular and circular patches, it was noted that the current distribution is seen at the edge of the circular patches (radiating elements) in 25.8 and 26.3 GHz, respectively, for Fig. 3 a and b. It is observed at 25.8 and 26.3 GHz; most of the power is radiated from the lower left and upper right circular patches of the antenna. In Fig. 3c, the current distribution was mainly concentrated around the edges of the inner circular slot within the lower rectangular patch adjacent to the feed line for a frequency of 27 GHz. Besides, the current was primarily distributed around the inner circular slot of the upper rectangular patch at 28.5 GHz as depicted in Fig. 3d.

Fig. 3
figure 3

Surface current distribution at different frequencies for the proposed design

Full size image

Thus, these focused surface currents on the edges of the circular slots inserted at the rectangular radiating elements make them more resourceful than the feed line. The essential effect of the constructive coupling of various antenna patches on the overall antenna performance is seen in Fig. 3 e and f. Due to this strong coupling, lots of energies are attached to various microstrip line elements, and the distributions of surface currents are then at 29.5 and 30.4 GHz. Nevertheless, the shift in some frequencies resonances is taken place due to coupling between various patches; these added resonance frequencies were later merged by the edges to achieve a multiband performance. Finally, the surface current is mainly concentrated on the edge of the feed line that leads practically to a surface current at 32 GHz as plotted in Fig. 3g.

4 Fabrication and Tests

Due to the insightful analysis, attractive current distribution features, and different frequency resonances, a truthful overview of the designed antenna performance is achieved. The antenna fabrication process was then done to validate the aforementioned simulation results and check its physical performance. The fabricated 5G antenna prototype (\({S}_{11}\)) and VSWR parameters were measured through 50 Ω port of the vector network analyzer (Rohde & Schwarz—Model ZVA 67). The test setup is shown in Fig. 4 where the antenna is connected via a coaxial cable to vector network analyzer (VNA) port 1 with the same specifications mentioned in [22]. Figure 5 shows the measuring antenna input reflection coefficient (\({S}_{11}\)) versus frequency in different assigned operating bandwidths. The (\({S}_{11}\)) antenna parameters are evaluated through contrasts between the simulated and the physical measurements. After the fabrication and measurement of the antenna, the improvement on (\({\mathrm{S}}_{11}\)) physical characteristics at a higher frequency was clearly shown. The frequency bands cantered practically at 25.1, 26.9, 28.4, 29.6 and 31 GHz have been allocated for 5G mobile communications by the proposed first dual-band 25–32 GHz as plotted in Fig. 5a, while the second bandwidth (40–60 GHz) comprises cantered frequencies at 40.3, 43, 46.5, 54, 57, and 58.6 GHz as presented in Fig. 5b. With regard to the comparison between the computed results and the measured one, there is a diffuse difference between the simulated and measured results as shown in Fig. 5. The differences are basically due to the fabrication tolerance, hand welding inaccuracy, as well as the mm-wave connector effect [23, 24]. Moreover, these uncertainties may be due to the measurement and fabrication process where the signal source from the network analyzer was not instantly frequency stabilized at both calibration and measurement times resulted in a phase error [25:27]. Moreover, the size and locations of the semicircular patches are optimized to obtain better 3D radiation characteristics and antenna gain.

Fig. 4
figure 4

Experimental setups of measuring the input reflection coefficient \({\mathrm{S}}_{11}\) by a VNA

Full size image
Fig. 5
figure 5

Measuring manufactured antenna input reflection coefficient (\({S}_{11}\)) values by a VNA

Full size image

The 3D radiation characteristics were simulated and measured at some selected frequencies as depicted in Table 2. The radiation efficiency of the planned antenna is with radiation power overlaps in a unidirectional pattern. A broadside direction was achieved in both planes either in measured or simulated 3D radiation patterns as plotted in Table 2. Moreover, the measured 2D gain values of the proposed antenna was demonstrated in Fig, 6, where a maximum gain of nearly 10 dBi at 25 and 28.5 GHz was achieved as shown in Fig. 6a. It could be noted that the proposed antenna maintains the same performance with gain values ranging from 7 up to 10.75 dBi during the operation in the higher frequencies as in Fig. 6b [22,23,24,25,26,27,28]. Furthermore, from Fig. 6, it is observed that the proposed antenna configuration has a wide-ranging variation in the gain magnitude in both 25–32 and 40–60 GHz frequency ranges. These significant variations in the magnitude are due to various regions of the antenna cause different edge impedance. Hence, the highest gain is achieved for better impedance matching, while the lowest gain is due to non-perfect impedance matching [29].

Fig. 6
figure 6

Simulated and measured 2D gain for the proposed antenna at both frequency bands

Full size image

The realized measured and simulated radiation efficiency of the proposed antenna is exhibited in Fig. 7. As shown in Fig. 7a, the proposed antenna has reasonable measured radiation efficiency reaching up to 85%, within the first frequency range 25–32 GHz. The simulated peak efficiency is reaching up to 84% during the 25–32 GHz bandwidth. In the second bandwidth 40–60, it can be observed that the manufactured antenna has a realized measured and simulated efficiency reaching up to 84% and 83.8% respectively. Therefore, the fluctuations of the gain and efficiency values are associated with the change in the impedance matching at different operating range frequencies [29].

Fig.7
figure 7

Simulated and measured plot of antenna radiation efficiency

Full size image

Finally, Table 3 shows a comparison between the proposed design and some lately reported results. As depicted from the table, the proposed antenna prototype achieved better (\({\mathrm{S}}_{11}\)), radiation characteristics, and gain than the previously designed structures. Compact size is also achieved within the entire bandwidth from 25 to 32 and 40 to 60 GHz. Thus, this prototype structure can be used as a strong microstrip antenna for cutting edge 5G technologies.

Table 3 A comparison between the proposed antenna and other published papers
Full size table

5 Conclusions

In this work, we design and fabricate a microstrip antenna operating at multiple-band operation covering the 25–32 and 40–60 GHz mm-wave frequency bands. These frequency bands have an implication in wireless applications, specifically 5G technologies. The design of the proposed antenna is discussed in detail. It is based on miniaturizing the overall antenna dimensions by using rectangular and circular radiator patches all together loaded on the same substrate. The resilient couplings between driven and parasitic patches offer multiband processing over the two different frequency ranges. Consequently, this configuration is fabricated, and its gain and radiation characteristics are measured experimentally. An enhancement of the radiation characteristics of the proposed antenna in the two operating bandwidths is thus achieved. The realized prototype shows multiband antenna performance at the specified frequency ranges with reflection coefficients of almost − 10 dB. The gain, radiation characteristics, and radiation efficiency of the offered antenna have been experimentally determined. The results show that the proposed antenna has a high gain and good directional radiation characteristics. The compactness of wide bandwidth, upstanding gain and radiation efficiency suggest that the proposed antenna is a prospective candidate for upcoming communication systems.